In recent decades ultrasonic imaging technology has played an increasing role in examining the internal structure of living organisms. The technology has applications in diagnosis of various medical ailments where it is useful to examine structural details in soft tissues within the body. Ultrasound imaging systems are advantageous for use in medical diagnosis as they are non-invasive, easy to use, and do not subject patients to the dangers of electromagnetic radiation. Instead of electromagnetic radiation, an ultrasound imaging system transmits sound waves of very high frequency (e.g., 2 MHz to 10 MHz) into the patient and processes echoes reflected from structures in the patient's body to form two dimensional or three dimensional images.
More recently, ultrasonic imaging systems have been able to detect blood flow as well as tissue amplitude along the axis of the interrogating ultrasonic wave. These blood velocities are related by the Doppler effect movement within the body and blood flow is information of high diagnostic significance for certain diseases. This and other ultrasound background information is disclosed in Zagzebski, Essentials of Ultrasound Physics (Mosby 1996), the contents of which are hereby incorporated by reference into the present disclosure.
Signal demodulation represents a key preliminary step in converting reflected ultrasound signals into a usable representation on an output display. FIGS. 1A and 1B show exemplary plots of ultrasonic waves as transmitted into the body and received from the body, respectively, by an ultrasound transducer (not shown). In a typical ultrasound system operating in pulse-echo mode, the transducer transmits a signal 100 comprising a series of time-limited modulated bursts as shown in FIG. 1A, and then receives a reflected signal 102 as shown in FIG. 1B. The plot 100 of FIG. 1A is of a generic voltage signal which, as known in the art, is converted into sound waves by piezoelectric devices in the transducer for transmission into the body. The plot 102 of FIG. 1B is likewise a generic voltage signal resulting from the conversion of reflected sound waves into voltages by the piezoelectric devices in the probe.
It is to be appreciated for purposes of the present disclosure that system specifics such as frequency ranges, pulse durations, and the like are given by way of example only and are not intended to limit the scope of the preferred embodiments. By way of example and not by way of limitation, typical parameters for real-time ultrasound imaging applications would include: a carrier frequency F.sub.c of 8 MHz (depending on the application, ultrasound implementations may use F.sub.c selected from the range 2 MHz-15 MHz); a burst repetition frequency BRF of 2000 Hz, i.e., 1 burst is sent every 500 .mu.s (real-time imaging systems may use a BRF selected from the range 2000 Hz-4000 Hz); and a burst duration BD of 1 .mu.s. As shown in FIG. 1B, the reflected signal x(t) in plot 102 can be expressed as a sinusoid at the carrier frequency F.sub.c modulated by an envelope signal A(t) and a time-varying phase term .phi.(t), as can be expressed by Equations (1)-(3): EQU x(t)=A(t)cos{.omega..sub.c t+.phi.(t)} {1} EQU .omega..sub.c =2.pi.F.sub.c {2} EQU x(t)=Re{A(t)e.sup.j.phi.(t) e.sup.j.omega.ct } {3}
Depending on the ultrasound implementation, it is the content of the envelope signal A(t) and/or the content of the phase signal .phi.(t) that is of interest in generating the ultrasound system display output. For example, in B-mode imaging it is the envelope signal A(t) that is of interest because A(t) is proportional to the reflective index of the target tissue. As another example, in Doppler mode imaging it is the phase signal .phi.(t) that is of interest because as the velocity of fluid flow in the target area is proportional to its first derivative d.phi./dt. Because the basic analytic signal A(t)e.sup.j.phi.(t) modulated by the carrier frequency F.sub.c in Equations (1)-(3) is usually found to contain frequency content in the several-megahertz range, which is not significantly less than the carrier frequency F.sub.c, the resultant signal x(t) is generally classified as a wideband signal.
FIG. 1C shows frequency spectra 104, 106, and 108 of the wideband signal x(t) corresponding to differing exemplary target depths of 1 cm, 10 cm, and 20 cm, respectively. The spectra 104, 106, and 108 correspond to the amplitude of the Fourier transform X(f) of the signal x(t) taken over small intervals of time around points corresponding to the expected arrival time of burst reflections from depths of 1 cm, 10 cm, and 20 cm, respectively. As illustrated by FIG. 1C, the frequency spectrum X(f) is time varying in its characteristics. As known in the art, this is due to a frequency and depth attenuation factor of about 0.6 dB/MHz-cm for a typical ultrasound system on a typical human target, which causes the spectrum to be increasingly skewed toward lower frequencies in the far field (i.e., at greater depths) than in the near field (i.e., at lesser depths). Accordingly, there are different half-power bandwidths W of the analytic signal A(t)e.sup.j.phi.(t) as a function of depth, and in general W.sub.depth=d1 &gt;W.sub.depth=d2 for d2&gt;d1. For purposes of clarity of disclosure, and not for purposes of limiting the scope of the preferred embodiments, the analytic signal A(t)e.sup.j.phi.(t) is characterized herein as having a generic upper frequency limit W/2, where W is a half-power bandwidth for a depth in the near field. It is to be understood, however, that in practical systems a modified value for the upper frequency limit may be used without departing from the scope of the preferred embodiments.
FIG. 2 shows a block diagram of a conventional ultrasound system 200 in accordance with prior art. Ultrasound system 200 comprises a transducer 202, a front end processor 204, a demodulator 206, and an amplitude detector and display apparatus 208. As described supra, transducer 202 transmits focused acoustic signals into the body and sends an analog response signal to front end processor 204. Front end processor 204 performs preliminary operations such as depth gain compensation and then transmits the result, commonly called the RF signal, to demodulator 206. For purposes of clarity and simplicity, the analog signal x(t), shown as plot 102 of FIG. 1B, is referred to herein as the RF signal, with depth gain compensation and/or other preliminary processing having already been performed by front end processor 204.
In the system of FIG. 2, the demodulator 206 performs processing steps directed to extracting the envelope signal A(t) and the phase signal .phi.(t) from the RF signal x(t), and then transmits the result to amplitude detector and display apparatus 208 for downstream processing and eventual image display. In a simplest form, for B-mode processing such a demodulator could simply comprise an analog envelope detector having a full-wave rectifier and a low-pass filter, as described in Carlson, Communication Systems: An Introduction to Signals and Noise in Electrical Communication (McGraw-Hill, 3.sup.rd ed. 1986), the contents of which are hereby incorporated by reference into the present disclosure. However, in most practical ultrasound systems in which both the amplitude signal A(t) will be desired for a B-mode processing mode, and where phase signal .phi.(t) will be desired for Doppler processing mode, a coherent detection scheme is used wherein demodulator 206 comprises a quadrature mixer. While older prior art implementations were analog in nature, most newer implementations are digital. For purposes of the present disclosure, and without loss of generality, it may be presumed that front end processor 204 incorporates an anti-aliasing filter followed by an analog to digital converter operating at a sampling frequency of F.sub.s.
FIG. 3 shows a block diagram of demodulator 206 in accordance with prior art. Demodulator 206 comprises a local oscillator 302, mixers 304 and 306, and low pass filters 308 and 310. In accordance with conventional digital signal processing principles, the sampling frequency F.sub.s is at least satisfying the Nyquist rate, which for the system of FIG. 3, 2(F.sub.c +W/2) is selected where, as described supra, F.sub.c is the ultrasound carrier frequency and W/2 is the upper frequency limit W/2 of the analytic signal A(t)e.sup.j.phi.(t). Upon sampling, the digitized RF signal x(kT) results, with the sampling period being T=1/F.sub.s, the signal x(kT) simply being cast herein as x(k). Local oscillator 302 is designed to generate sinusoids at a mixing frequency F.sub.x that are in quadrature phase with each other, e.g., cos(2.pi.F.sub.x k) and sin(2.pi.F.sub.x k). Mixers 304 and 306 multiply the digitized RF signal x(k) with the respective quadrature-phase signals from the local oscillator 302, and the products are sent to low-pass filters 308 and 310, respectively. Low-pass filters 308 and 310 are designed to have sharp rolloffs at the upper frequency limit W/2 of the analytic signal A(t)e.sup.j.phi.(t).
As known in the art, for Doppler processing the mixing frequency F.sub.x is fixed at a frequency identical to the carrier frequency F.sub.c. Using relationships known in the art and as described generally in Carlson, supra, in such case the demodulator 206 generates the signals I(k) and Q(k), where I(k) and Q(k) are related to the amplitude signal A(t) (expressed herein as A(k) in the digital domain) and phase signal .phi.(t) (expressed herein as .phi.(k) in the digital domain) as shown in Equations (4)-(8) below: EQU A(k)e.sup.j.phi.(k) =I(k)+jQ(k) {4} EQU I(k)=A(k)cos.phi.(k) {5} EQU Q(k)=A(k)sin.phi.(k) {6} EQU A(k)={I.sup.2 (k)+Q.sup.2 (k)} {7} EQU .phi.(k)=tan.sup.-1 {Q(k)/I(k)} {8}
For B-mode processing, where it is desirable to obtain amplitude A(k) and where detection of phase .phi.(k) is not important, the mixing frequency F.sub.x is selected to be time-varying for increasing the signal-to-noise ratio. In particular, the mixing frequency F.sub.x is selected to correspond to an instantaneous center frequency of the signal x(t) which, as shown in FIG. 1C, becomes increasingly lower than the carrier frequency F.sub.c as the field depth increases. In such case, the oscillator 302 is a swept frequency oscillator, with the mixer frequency F.sub.x varying from a maximum value to a minimum value over each burst period BP. As known in the art, using a swept oscillator for the mixing frequency F.sub.x provides better signal-to-noise performance and therefore better detection of A(k) as compared to using a mixing frequency F.sub.x fixed at the carrier frequency F.sub.c. However, it can also become impractical to recover the phase signal .phi.(k) using this method.
By way of nonlimiting example, and further to the example of FIG. 1C, for B-mode processing the mixing frequency F.sub.x may sweep from a maximum of 7.5 MHz in the near field to 5.0 MHz in the far field. It can be shown for such B-mode processing that the amplitude signal A(k) can still be computed from the demodulator 206 outputs I(k) and Q(k) from Equation (7) above while, as discussed supra, the phase signal .phi.(k) is generally not computed.
FIG. 4, comprising parts 4-1 and 4-2 shows Fourier spectrum plots corresponding to the operation of prior art demodulator 206 of FIG. 3. FIG. 4 shows a first plot 402 of the Fourier transform X(f) of the RF signal x(k), a second plot 404 of the Fourier transform of the local oscillator output cos(.omega..sub.x kT), a third plot 406 of the Fourier transform of the mixer 304 output, a fourth plot 410 of the Fourier transform of the local oscillator output sin(.omega..sub.x kT) as multiplied by j, and a fifth plot 412 of the Fourier transform of the mixer 306 output as multiplied by j. Note that Fourier transform plots 402, 406, and 412 repeat at intervals of F.sub.s as they represent digital signals at a sampling frequency of F.sub.s, and only the frequency range of -F.sub.s to F.sub.s is shown.
Several problems, however, arise in ultrasound systems using the prior art wideband demodulator scheme of FIGS. 2 and 3. As shown on spectrum plots 406 and 412, the low-pass filters 308 and 310 must each have a frequency characteristic similar to the filter characteristic 408 for appropriate mirror rejection, i.e. cancellation of the undesired mirror portion of the mixed signals that occur at the output of mixers 304 and 306, respectively. As shown in FIG. 4, filter characteristic 408 of low-pass filters 308 and 310 is required to have a very sharp roll-off for proper construction of the signals I(k) and Q(k), respectively. In FIG. 4, for example, it is shown that the distance between the desired signal and its undesired mirror signal is approximately (F.sub.s -2F.sub.c -W) if F.sub.s &lt;2(F.sub.c +F.sub.x), and approximately 2(F.sub.x -W) if F.sub.s &gt;=2(F.sub.c +F.sub.x). Using the exemplary parameters of FIG. 4, this distance is only 32-2(8)-6.4=9.6 Mhz. Implementing this sharp roll-off, however, can create large group delay distortion at the filter band edge that degrade images produced from these signals and cause ringing. Additionally, using prior art ultrasound sampling and filter frequencies, the computations for implementing the low pass filters 308 and 310 are complex and time consuming, requiring general purpose digital signal processors of a very high order and very high power. Disadvantageously, the hardware needed to implement the prior art demodulator suffers from increased size, heat dissipation requirements, cost, and complexity due to the need for the sharp roll-off characteristic of low pass filters 308 and 310.
An additional problem with prior art demodulators as in FIG. 3 arises in the context of B-mode processing, which usually incorporates a swept mixing frequency F.sub.x at local oscillator 304 as described supra. Because the cutoff frequency of each of the low pass filters 308 and 310 is fixed at a single value, the mirror cancellation performed by low pass filters 308 and 310 is inherently better for certain values of mixing frequency F.sub.x than others. A undesirable design choice is forced upon the ultrasound hardware designer. Given this dilemma, the design choice is usually made to select the cutoff frequency of the filters 308 and 310 such that near field reflections receive better mirror-canceling performance than far-field reflections. As a result of this design choice, there is inferior demodulation of far-field reflections as compared to near-field reflections.
One attempted prior solution to the above problems is described in U.S. Pat. No. 5,482,044, the contents of which are hereby incorporated by reference into the present application. The method described therein, however, which includes the use of a Hilbert transform, suffers from practical implementation difficulties and complexities, including problems related to the fact that the Hilbert transform is a noncausal signal which is not realizable in a physical system and can only be approximated.
A further problem with the prior art ultrasound information processing system of FIGS. 2 and 3, as well as U.S. Pat. No. 5,482,044, supra, is found in the context of harmonic imaging. As known in the art, in harmonic imaging mode an ultrasound information processing system attempts to recover second harmonic information near 2F.sub.c. The above cited prior art systems, however, provide no easy digital solution to the recovery of the second harmonic information, and in field implementations there is usually an analog high pass filter incorporated into the front end processor 204 that removes, in the analog domain, the information near F.sub.c. The requirement to include an adjustable analog filter in the front end processor 204 with sufficient high pass characteristics presents problems of cost and complexity that it would be to desirable to avoid.
Accordingly, it would be desirable to provide an ultrasound demodulator having an improved output with better mirror canceling, less group delay distortion, and less ringing.
It would be further desirable to provide an ultrasound demodulator having an improved output while also being easier to implement in hardware, using lower cost, off-the-shelf components with lesser footprint and heat dissipation requirements.
It would be still further desirable to provide an ultrasound demodulator that, in amplitude-only implementations having swept-frequency mixing at a local oscillator, produces improved output in the far field as well as the near field.
It would be still further desirable to provide an ultrasound information processing system having the capability of performing harmonic imaging using low-cost, off-the-shelf components that implement a digital technique incorporated into the ultrasound demodulator hardware, and not requiring an analog high pass filter at a front end processor of the ultrasound information processing system.